There is known a high-frequency power source configured to convert direct current (DC) from a DC power source into high-frequency alternating current (AC), according to switching operation. As one of such high-frequency power sources, a class-D high-frequency power source is known, which is based on the class-D amplifier circuit (Class D: IEC international standard IEC60268-3, 4 classes of operation).
The class-D high-frequency power source incorporates RF power amplifying elements to form a full-bridge circuit or a half-bridge circuit. Then, employing this RF power amplifying element as a switching element, a switching operation is performed using RF gate signals with constant duty cycle, thereby converting direct current from the DC power source into high-frequency alternating current, and supplying to a load, thus obtained high-frequency alternating current serving as high-frequency traveling wave power. The class-D high-frequency power source performs output adjustment, either in the pulse drive mode or in the continuous drive mode. In the pulse drive mode, the RF gate signal causes the RF power amplifying element to perform the switching operation, being provided with ON intervals and OFF intervals alternately; in the ON interval, there is an RF output, and in the OFF interval there is no RF output without performing the switching operation. Then, by changing the duty being a time ratio between the ON interval and the OFF interval, output power of the RF output is adjusted. This duty between the ON interval and the OFF interval may be controlled by the duty between the ON interval and the OFF interval of a pulse control signal. Here, “RF” indicates high frequency.
FIG. 11 illustrates a configuration example of a class-D amplifier circuit for RF band, according to a conventionally known bridge circuit configuration. The example here shows a configuration that switching elements being MOSFETs (Metal-Oxide-Semiconductor Field-Effect Transistors) constitute the full-bridge circuit. The class-D amplifier circuit converts direct current into high-frequency alternating current which is supplied to the load. Here, the direct current is supplied from the DC power supply part, via the wiring inductance La and lead-wire inductance Lp of the lead wire provided on the class-D amplifier circuit package.
In the low RF range where switching frequency is not high, the bypass capacitor C2 is connected between the lead wires outside the package, so as to suppress the surge voltage that is generated at ON/OFF time of the MOSFET.
The surge voltage is suppressed by the bypass capacitor C2 according to the following conditions; in the low RF range, drain-source voltage rise time during the ON time of the MOSFET and the drain-source voltage fall time during the OFF time of the MOSFET are relatively slow going, with small current variation (di/dt) in the MOSFET, and therefore, switching loss in the low RF range of the MOSFET is larger than the switching loss in the high RF band.
In addition, the bypass capacitor C2 is configured to be large, whereby the switching loss in the MOSFET serves as damping resistance, and producing an effect of suppressing resonance phenomenon caused by the wiring inductance La.
FIG. 12A illustrates an equivalent circuit of the class-D amplifier circuit as shown in FIG. 11, when it is switched from ON to OFF. FIG. 12B schematically illustrates the drain-source voltage of the class-D amplifier circuit for the low RF range. In the equivalent circuit, “r” represents the switching loss, “Lp” represents the lead-wire inductance of the package, and “Cd” represents capacitance in the package.
In the low RF range, the characteristic equation for the class-D amplifier circuit may be quadratic, and the attenuation coefficient ζ is expressed as; ζ=r/2·(2Cd/Ld)1/2. The switching loss r works as the damping resistance, and attenuates the drain-source voltage oscillation at the time of ON/OFF switching of the MOSFET as shown in FIG. 12B, assuming the attenuation coefficient ζ as a value larger than 1.
On the other hand, since the MOSFET used for the high RF range where the switching frequency is high operates at high speed, the current variation (di/dt) in the MOSFET becomes large. FIG. 13 illustrates an example of the class-D amplifier circuit for the high RF range. It is to be noted that the circuit surrounded by the broken line in FIG. 13 indicates a snubber circuit which will be described later. FIG. 14A illustrates an equivalent circuit of the class-D amplifier circuit (circuit configuration excluding the snubber circuit) for the high RF range as shown in FIG. 13, when it is switched from ON to OFF. FIG. 14B schematically illustrates the drain-source voltage of the class-D amplifier circuit (circuit configuration excluding the snubber circuit) for the high RF range. The reference “La” in the equivalent circuit represents the wiring inductance outside the package, “Lp” represents the lead-wire inductance of the package, and “Cd” represents the capacitance in the package.
In the high RF range, since the wiring inductance Ld and the capacitance Cd within the package become considerable values, it is not possible for the bypass capacitor C2 provided outside the package to suppress the drain-source surge voltage in the MOSFET.
As illustrated by the equivalent circuit in FIG. 14A, since the switching loss in the high-speed MOSFET is small and the switching loss r does not exist in the class-D amplifier circuit for the high RF range, it does not work as the damping resistance. Therefore, as indicated by C and D in FIG. 14B, it is not possible to produce the effect to suppress oscillation of the drain-source voltage, and thus resonance phenomenon occurs continuously. The reference C in the figure indicates the voltage oscillation state at the time when the switching element is switched from OFF to ON, and the surge voltage is generated. The reference D in the figure indicates the voltage oscillation state at the time when the switching element is switched from ON to OFF. At any of the switching times, thus generated oscillation state may continue until the next switching time.
As shown in FIG. 14B, when surge voltage is generated between drain and source in the MOSFET, there may occur problems that efficiency may be lowered in converting DC to AC, and in addition, excess surge voltage may cause element destruction.
The bypass capacitor C2 connected to the outside the package is not able to absorb the surge energy in the lead-wire inductance Lp and in the wiring inductance Ld in the package, so as to prevent a flow of the surge energy into the MOSFET.
As a measure taken for the high RF range, in addition to providing an isolator, a 3 dB coupler, or the like, on the load side, there is known a configuration that a CR snubber circuit is connected between the supply lines, the CR snubber circuit formed by a series circuit of Csu and Rsu, aiming at suppressing natural oscillation caused by the inductance outside the package.
In addition, a bypass capacitor is connected in parallel, to the series circuit of switching elements constituting the half-bridge circuit within the package, so as to avoid reduction of the supply voltage when current pulses occur at the time of switching, and deter damage on the switching elements due to the current peak (Patent Document 1).
FIG. 15 illustrates a configuration example that the bypass capacitor is connected in parallel to the half-bridge circuit. The bypass capacitor C3 is configured as connected in parallel to the series circuit of the switching elements that constitute the half-bridge circuit.
In the configuration that provides the bypass capacitor within the package of the class-D amplifier, there is a problem that the switching elements and the bypass capacitor form a current path, and the line inductance within the current path and the output capacitance from the switching elements may form an oscillation circuit. Therefore, it is pointed out that application of the class-D amplifier to plasma processes is inadequate (paragraph [0012] in the Patent Document 1). The Patent Document 1 suggests that the length of the current path flowing in the bypass capacitor is restricted, so as to make the inductance smaller and shift oscillation frequencies to higher frequencies, thereby facilitating damping (paragraph in the Patent Document 1).